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19-2622; Rev 0; 10/02
1MHz, All-Ceramic, 2.6V to 5.5V Input, 1.5A PWM Step-Down DC-to-DC Regulators
General Description
The MAX1951/MAX1952 high-efficiency, DC-to-DC step-down switching regulators deliver up to 1.5A of output current. The devices operate from an input voltage range of 2.6V to 5.5V and provide an output voltage from 0.8V to VIN, making the MAX1951/MAX1952 ideal for on-board postregulation applications. The MAX1951 total output error is less than 1% over load, line, and temperature. The MAX1951/MAX1952 operate at a fixed frequency of 1MHz with an efficiency of up to 94%. The high operating frequency minimizes the size of external components. Internal soft-start control circuitry reduces inrush current. Short-circuit and thermal-overload protection improve design reliability. The MAX1951 provides an adjustable output from 0.8V to VIN, whereas the MAX1952 has a preset output of 1.8V. Both devices are available in a space-saving 8-pin SO package. o Compact 0.385in2 Circuit Footprint o 10F Ceramic Input and Output Capacitors, 2H Inductor o Efficiency Up to 94% o 1% Output Accuracy Over Load, Line, and Temperature (MAX1951) o Guaranteed 1.5A Output Current at +85C o Operate from 2.6V to 5.5V Supply o Adjustable Output from 0.8V to VIN (MAX1951) o Preset Output of 1.8V (1.5% Accuracy) (MAX1952) o Internal Digital Soft-Soft o Short-Circuit and Thermal-Overload Protection
Features
MAX1951/MAX1952
Applications
ASIC/DSP/P/FPGA Core and I/O Voltages Set-Top Boxes Cellular Base Stations Networking and Telecommunications
PART MAX1951ESA
Ordering Information
TEMP RANGE PINPACKAGE OUTPUT Adj 0.8V to VIN Fixed 1.8V
-40C to +80C 8 SO
MAX1952ESA-18 -40C to +80C 8 SO
Pin Configuration
Typical Operating Circuit
INPUT 2.6V TO 5.5V IN LX OUTPUT 0.8V TO VIN, 1.5A
TOP VIEW
VCC 1 REF GND 2 3
8 7
IN LX PGND COMP ON OFF
MAX1951
VCC FB
MAX1951 MAX1952
6 5
FB 4
COMP PGND
REF GND
SO
OPTIONAL
________________________________________________________________ Maxim Integrated Products
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1MHz, All-Ceramic, 2.6V to 5.5V Input, 1.5A PWM Step-Down DC-to-DC Regulators MAX1951/MAX1952
ABSOLUTE MAXIMUM RATINGS
IN, VCC to GND ........................................................-0.3V to +6V COMP, FB, REF to GND .............................-0.3V to (VCC + 0.3V) LX to Current (Note 1).........................................................4.5A PGND to GND .............................................Internally Connected Continuous Power Dissipation (TA = +85C) 8-Pin SO (derate 12.2mW/C above +70C)................976mW Operating Temperature Range MAX195_ ESA..................................................-40C to +85C Junction Temperature Range ............................-40C to +150C Storage Temperature Range .............................-65C to +150C Lead Temperature (soldering, 10s) .................................+300C
Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
Note 1: LX has internal clamp diodes to PGND and IN. Applications that forward bias these diodes should take care not to exceed the IC's package power dissipation limits.
ELECTRICAL CHARACTERISTICS
(VIN = VCC = 3.3V, PGND = GND, FB in regulation, CREF = 0.1F, TA = 0C to +85C, unless otherwise noted. Typical values are at TA = +25C.)
PARAMETER IN AND VCC IN Voltage Range Supply Current Shutdown Current VCC Undervoltage Lockout Threshold REF REF Voltage REF Load Regulation REF Line Regulation REF Shutdown Resistance COMP COMP Transconductance COMP Clamp Voltage, Low COMP Clamp Voltage, High COMP Shutdown Resistance COMP Shutdown Threshold COMP Startup Current FB Output Voltage Range (MAX1951) FB Regulation Voltage (Error Amp Only) FB Input Resistance FB Input Bias Current When using external feedback resistors to drive FB VCOMP = 1V to 2V, IOUT = 0 to 1.5A VIN = 2.6V to 5.5V VIN = 2.8V to 5.5V MAX1951 MAX1952 MAX1952 MAX1951 0.8 0.787 1.773 13 -0.1 0.795 1.8 18 VIN 0.803 1.827 28 +0.1 V V k A From FB to COMP, VCOMP = 1.25V VIN = 2.6V to 5.5V, VFB = 1.3V VIN = 2.6V to 5.5V, VFB = 1.1V From COMP to GND, VIN = 2V When LX starts/stops switching COMP = GND COMP rising COMP falling 0.17 15 MAX1951 MAX1952 40 26.7 0.6 1.97 60 40 1 2.15 15 0.6 0.4 25 40 80 53.3 1.2 2.28 30 1 S V V V A IREF = 0, VIN = 2.6V to 5.5V IREF = 0 to 40A, VIN = 2.6V to 5.5V IREF = 20A, VIN = 2.6V to 5.5V From REF to GND, COMP = GND 1.96 2 0.01 0.01 12 2.03 0.2 0.4 22 V % % Switching with no load, LX floating COMP = GND When LX starts/stops switching VCC rising VCC falling 2 VIN = 5.5V 2.6 6 0.5 2.35 2.25 5.5 10 1.0 2.5 V mA mA V CONDITIONS MIN TYP MAX UNITS
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1MHz, All-Ceramic, 2.6V to 5.5V Input, 1.5A PWM Step-Down DC-to-DC Regulators
ELECTRICAL CHARACTERISTICS (continued)
(VIN = VCC = 3.3V, PGND = GND, FB in regulation, CREF = 0.1F, TA = 0C to +85C, unless otherwise noted. Typical values are at TA = +25C.)
PARAMETER LX VIN = 5V LX On-Resistance, PMOS VIN = 3.3V VIN = 2.6V VIN = 5V LX On-Resistance, NMOS LX Current-Sense Transimpedance LX Current-Limit Threshold LX Leakage Current LX Switching Frequency LX Maximum Duty Cycle LX Minimum Duty Cycle VIN = 3.3V VIN = 2.6V From LX to COMP, VIN = 2.6V to 5.5V Duty cycle = 100%, VIN = 2.6V to 5.5V VIN = 5.5V VIN = 2.6V to 5.5V VCOMP = 1.5V, LX = high-Z, VIN = 2.6V to 5.5V VCOMP = 1V, VIN = 2.6V to 5.5V TJ rising TJ falling High side Low side VLX = 5.5V LX = GND -10 0.85 100 15 160 145 1 1.1 0.16 2.4 116 140 163 93 106 116 0.24 3.1 -0.6 10 0.35 4.5 A A MHz % % 206 m 266 m CONDITIONS MIN TYP MAX UNITS
MAX1951/MAX1952
THERMAL CHARACTERISTICS Thermal-Shutdown Threshold When LX starts/stops switching C
ELECTRICAL CHARACTERISTICS
(VIN = VCC = 3.3V, PGND = GND, FB in regulation, CREF = 0.1F, TA = -40C to +85C, unless otherwise noted.) (Note 2)
PARAMETER IN AND VCC IN Voltage Range Supply Current Shutdown Current VCC Undervoltage Lockout Threshold REF REF Voltage REF Load Regulation REF Line Regulation REF Shutdown Resistance IREF = 0, VIN = 2.6V to 5.5V IREF = 0 to 40A, VIN = 2.6V to 5.5V IREF = 20A, VIN = 2.6V to 5.5V From REF to GND, COMP = GND 1.95 2.03 0.2 0.4 22 V % % Switching with no load, VIN = 5.5V COMP = GND When LX starts/stops switching VCC rising VCC falling 1.95 2.6 5.5 10 1 2.5 V mA mA V CONDITIONS MIN TYP MAX UNITS
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1MHz, All-Ceramic, 2.6V to 5.5V Input, 1.5A PWM Step-Down DC-to-DC Regulators MAX1951/MAX1952
ELECTRICAL CHARACTERISTICS (continued)
(VIN = VCC = 3.3V, PGND = GND, FB in regulation, CREF = 0.1F, TA = -40C to +85C, unless otherwise noted.) (Note 2)
PARAMETER COMP COMP Transconductance COMP Clamp Voltage, Low COMP Clamp Voltage, High COMP Shutdown Resistance COMP Shutdown Threshold COMP Startup Current FB Output Voltage Range (MAX1951) FB Regulation Voltage (Error Amp Only) FB Input Resistance FB Input Bias Current LX LX On-Resistance, PMOS LX On-Resistance, NMOS LX Current Sense LX Current-Limit Threshold LX Leakage Current LX Switching Frequency LX Maximum Duty Cycle From LX to COMP, VIN = 2.6V to 5.5V Duty cycle = 100%, VIN = 2.6V to 5.5V, high side VIN = 5.5V VIN = 2.6V to 5.5V VCOMP = 1.5V, LX = Hi-Z, VIN = 2.6V to 5.5V VLX = 5.5V LX = GND -10 0.8 100 1.1 0.16 2.4 266 206 0.35 4.5 10 m m A A MHz % When using external feedback resistors to drive FB VCOMP = 1V to 2V, VIN = 2.6V to 5.5V From FB to GND MAX1951 MAX1952 MAX1952 MAX1951 0.8 0.783 1.764 10 -0.1 VIN 0.807 1.836 30 +0.1 V V k A From FB to COMP, VCOMP = 1.25V VIN = 2.6V to 5.5V, VFB = 1.3V VIN = 2.6V to 5.5V, VFB = 1.1V From COMP to GND, VIN = 2V When LX starts/stops switching COMP = GND COMP rising COMP falling 0.17 14 40 MAX1951 MAX1952 40 26.7 0.6 1.97 80 53.3 1.2 2.28 30 1.2 S V V V A CONDITIONS MIN TYP MAX UNITS
Note 2: Specifications to -40C are guaranteed by design and not production tested. Note 3: The LX output is designed to provide 2A RMS current.
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1MHz, All-Ceramic, 2.6V to 5.5V Input, 1.5A PWM Step-Down DC-to-DC Regulators MAX1951/MAX1952
Typical Operating Characteristics
(Typical values are at VIN = VCC = 5V, VOUT = 1.5V, IOUT = 1.5A, and TA = +25C, unless otherwise noted. See Figure 2.)
EFFICIENCY vs. LOAD CURRENT (VCC = VIN = 5V)
MAX 1951 toc01
EFFICIENCY vs. LOAD CURRENT (VCC = VIN = 3.3V)
MAX 1951 toc02
REF VOLTAGE vs. REF OUTPUT CURRENT
MAX1951 toc03
100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 0 10
VOUT = 3.3V
100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 0
VOUT = 2.5V
1.995 1.994 REF VOLTAGE (V) 1.993 1.992 1.991 1.990 1.989 TA = -40C TA = +85C TA = +25C
VOUT = 2.5V VOUT = 1.5V
VOUT = 1.8V VOUT = 1.5V
VOUT = 0.8V
VOUT = 0.8V
100
1000
10,000
10
100
1000
10,000
0
5
10
15
20
25
30
35
40
LOAD CURRENT (mA)
LOAD CURRENT (mA)
REF OUTPUT CURRENT (A)
SWITCHING FREQUENCY vs. INPUT VOLTAGE
MAX1951 toc04
OUTPUT VOLTAGE DEVIATION vs. LOAD CURRENT
6 5 4 3 2 1 0 -1 -2 -3 -4 -5 -6 0 VOUT = 2.5V VOUT = 3.3V
MAX1951 toc05
1.20 1.15 SWITCHING FREQUENCY (MHz) 1.10 1.05 1.00 0.95 0.90 0.85 0.80 2.6 3.1 3.6 4.1 4.6 5.1 TA = -40C TA = +25C TA = +85C
OUTPUT VOLTAGE DEVIATION (mV)
VOUT = 0.8V
VOUT = 1.8V
5.6
0.4
0.8 LOAD CURRENT (A)
1.2
1.6
INPUT VOLTAGE (V)
LOAD TRANSIENT RESPONSE
MAX1951 toc06
LOAD TRANSIENT RESPONSE
MAX1951 toc07
OUTPUT VOLTAGE: 100mV/div, AC-COUPLED
OUTPUT VOLTAGE: 100mV/div, AC-COUPLED
OUTPUT CURRENT: 0.5A/div VIN = 5V VOUT = 2.5V IOUT = 0.5 TO 1A 40s/div
OUTPUT CURRENT: 0.5A/div VIN = 3.3V VOUT = 1.5V IOUT = 0.5 TO 1A 40s/div
0
0
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1MHz, All Ceramic, 2.6V to 5.5V Input, 1.5A PWM Step-Down DC-to-DC Regulators MAX1951/MAX1952
Typical Operating Characteristics (continued)
(Typical values are at VIN = VCC = 5V, VOUT = 1.5V, IOUT = 1.5A, and TA = +25C, unless otherwise noted. See Figure 2.)
SWITCHING WAVEFORMS
MAX1951 toc08
SOFT-START WAVEFORMS
MAX1951 toc09
INDUCTOR CURRENT 1A/div
VCOMP 2V/div
0 0 VLX 5V/div OUTPUT VOLTAGE 10mV/div, AC-COUPLED OUTPUT VOLTAGE 1V/div VIN = VCC = 3.3V VOUT = 2.5V ILOAD = 1.5A 1ms/div
VIN = 3.3V VOUT = 1.8V ILOAD = 1.5A 200ns/div
SOFT-START WAVEFORMS
MAX1951 toc10
SHUTDOWN WAVEFORMS
MAX1951 toc11
0 VCOMP 2V/div
VCOMP 2V/div
0 OUTPUT VOLTAGE 0.5V/div VIN = VCC = 3.3V VOUT = 0.8V 0 1ms/div 20s/div VIN = VCC = 3.3V VOUT = 2.5V ILOAD = 1.5A
VLX 5V/div
OUTPUT VOLTAGE 1V/div
SHUTDOWN CURRENT vs. INPUT VOLTAGE
0.9 SHUTDOWN CURRENT (mA) 0.8 0.7 0.6 0.5 0.4 0.3 0.2 0.1 0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 INPUT VOLTAGE (V)
MAX1951 toc12
1.0
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1MHz, All-Ceramic, 2.6V to 5.5V Input, 1.5A PWM Step-Down DC-to-DC Regulators
Pin Description
PIN NAME 1 2 3 VCC REF GND FUNCTION Supply Voltage. Bypass with 0.1F capacitor to ground and 10 resistor to IN. Reference Bypass. Bypass with 0.1F capacitor to ground. Ground Feedback Input. Connect to the output to regulate using the internal feedback resistor string (MAX1952). Connect an external resistordivider from the output to FB and GND to set the output to a voltage between 0.8V and VIN (MAX1951).
Controller Block Function
The MAX1951/MAX1952 step-down converters use a PWM current-mode control scheme. An open-loop comparator compares the integrated voltage-feedback signal against the sum of the amplified current-sense signal and the slope compensation ramp. At each rising edge of the internal clock, the internal high-side MOSFET turns on until the PWM comparator trips. During this on-time, current ramps up through the inductor, sourcing current to the output and storing energy in the inductor. The currentmode feedback system regulates the peak inductor current as a function of the output voltage error signal. Since the average inductor current is nearly the same as the peak inductor current (<30% ripple current), the circuit acts as a switch-mode transconductance amplifier. To preserve inner-loop stability and eliminate inductor staircasing, a slope-compensation ramp is summed into the main PWM comparator. During the second half of the cycle, the internal high-side P-channel MOSFET turns off, and the internal low-side N-channel MOSFET turns on. The inductor releases the stored energy as its current ramps down while still providing current to the output. The output capacitor stores charge when the inductor current exceeds the load current, and discharges when the inductor current is lower, smoothing the voltage across the load. Under overload conditions, when the inductor current exceeds the current limit (see the Current Limit section), the high-side MOSFET does not turn on at the rising edge of the clock and the low-side MOSFET remains on to let the inductor current ramp down.
MAX1951/MAX1952
4
FB
5
Regulator Compensation. Connect series RC network to GND. Pull COMP below 0.17V to COMP shut down the regulator. COMP = GND when VIN is less than 2.25V (see the Compensation and Shutdown Mode section) Power Ground. Internally connected to GND. PGND Keep power ground and signal ground planes separate. LX Inductor Connection. Connect an inductor between LX and the regulator output. Power-Supply Voltage. Input voltage range from 2.6V to 5.5V. Bypass with a 10F (min) ceramic capacitor to GND and a 10 resistor to VCC.
6
7
8
IN
Current Sense
An internal current-sense amplifier produces a current signal proportional to the voltage generated by the high-side MOSFET on-resistance and the inductor current (RDS(ON) x ILX). The amplified current-sense signal and the internal slope compensation signal are summed together into the comparator's inverting input. The PWM comparator turns off the internal high-side MOSFET when this sum exceeds the output from the voltage-error amplifier.
Detailed Description
The MAX1951/MAX1952 high-efficiency switching regulators are small, simple, DC-to-DC step-down converters capable of delivering up to 1.5A of output current. The devices operate in pulse-width modulation (PWM) at a fixed frequency of 1MHz from a 2.6V to 5.5V input voltage and provide an output voltage from 0.8V to VIN, making the MAX1951/MAX1952 ideal for on-board postregulation applications. The high switching frequency allows for the use of smaller external components, and internal synchronous rectifiers improve efficiency and eliminate the typical Schottky free-wheeling diode. Using the onresistance of the internal high-side MOSFET to sense switching currents eliminates current-sense resistors, further improving efficiency and cost. The MAX1951 total output error over load, line, and temperature (0C to +85C) is less than 1%.
Current Limit
The internal high-side MOSFET has a current limit of 3.1A (typ). If the current flowing out of LX exceeds this limit, the high-side MOSFET turns off and the synchronous rectifier turns on. This lowers the duty cycle and causes the output voltage to droop until the current limit is no longer exceeded. A synchronous rectifier current limit of -0.6A (typ) protects the device from current flowing into LX. If the negative current limit is exceeded, the synchronous rectifier turns off, forcing the inductor current to flow
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1MHz, All-Ceramic, 2.6V to 5.5V Input, 1.5A PWM Step-Down DC-to-DC Regulators MAX1951/MAX1952
POSITIVE AND NEGATIVE CURRENT LIMITS VCC OSC CLOCK CURRENT SENSE PWM CONTROL RAMP GEN SLOPE COMP CLAMP COMP SOFT-START/ UVLO BANDGAP REF 1.25V REF 2V ERROR SIGNAL THERMAL SHUTDOWN gm PGND FB LX IN
DAC
REF
MAX1951
GND
Figure 1. Functional Diagram
through the high-side MOSFET body diode, back to the input, until the beginning of the next cycle or until the inductor current drops to zero. The MAX1951/MAX1952 utilize a pulse-skip mode to prevent overheating during short-circuit output conditions. The device enters pulseskip mode when the FB voltage drops below 300mV, limiting the current to 3A (typ) and reducing power dissipation. Normal operation resumes upon removal of the short-circuit condition.
Undervoltage Lockout
If V CC drops below 2.25V, the UVLO circuit inhibits switching. Once V CC rises above 2.35V, the UVLO clears, and the soft-start sequence activates.
Compensation and Shutdown Mode
The output of the internal transconductance voltage error amplifier connects to COMP. The normal operation voltage for COMP is 1V to 2.2V. To shut down the MAX1951/MAX1952, use an NPN bipolar junction transistor or a very low output capacitance open-drain MOSFET to pull COMP to GND. Shutdown mode causes the internal MOSFETs to stop switching, forces LX to a high-impedance state, and shorts REF to GND. Release COMP to exit shutdown and initiate the softstart sequence.
VCC Decoupling
Due to the high switching frequency and tight output tolerance (1%), decouple VCC with a 0.1F capacitor connected from VCC to GND, and a 10 resistor connected from VCC to IN. Place the capacitor as close to VCC as possible.
Soft-Start
The MAX1951/MAX1952 employ digital soft-start circuitry to reduce supply inrush current during startup conditions. When the device exits undervoltage lockout (UVLO), shutdown mode, or restarts following a thermal-overload event, or the external pulldown on COMP is released, the digital soft-start circuitry slowly ramps up the voltages at REF and FB (see the Soft-Start Waveforms in the Typical Operating Characteristics).
Thermal-Overload Protection
Thermal-overload protection limits total power dissipation in the device. When the junction temperature exceeds TJ = +160C, a thermal sensor forces the device into shutdown, allowing the die to cool. The thermal sensor turns the device on again after the junction temperature cools by 15C, resulting in a pulsed output during continuous overload conditions. Following a thermal-shutdown condition, the soft-start sequence begins.
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1MHz, All-Ceramic, 2.6V to 5.5V Input, 1.5A PWM Step-Down DC-to-DC Regulators
Design Procedure
Output Voltage Selection: Adjustable (MAX1951) or Preset (MAX1952)
The MAX1951 provides an adjustable output voltage between 0.8V and VIN. Connect FB to output for 0.8V output. To set the output voltage of the MAX1951 to a voltage greater than VFB (0.8V typ), connect the output to FB and GND using a resistive divider, as shown in Figure 2a. Choose R2 between 2k and 20k, and set R3 according to the following equation: R3 = R2 x [(VOUT / VFB) - 1] The MAX1951 PWM circuitry is capable of a stable minimum duty cycle of 18%. This limits the minimum output voltage that can be generated to 0.18 VIN. Instability may result for VIN/VOUT ratios below 0.18. The MAX1952 provides a preset output voltage. Connect the output to FB, as shown in Figure 2b. For duty ratios less than 0.5, the input capacitor RMS current is higher than the calculated current. Therefore, use a +20% margin when calculating the RMS current at lower duty cycles. Use ceramic capacitors for their low ESR, equivalent series inductance (ESL), and lower cost. Choose a capacitor that exhibits less than 10C temperature rise at the maximum operating RMS current for optimum long-term reliability. After determining the input capacitor, check the input ripple voltage due to capacitor discharge when the high-side MOSFET turns on. Calculate the input ripple voltage as follows: VIN_RIPPLE = (IOUT x VOUT) / (fSW x VIN x CIN) Keep the input ripple voltage less than 3% of the input voltage.
MAX1951/MAX1952
Output Capacitor Design
The key selection parameters for the output capacitor are capacitance, ESR, ESL, and the voltage rating requirements. These affect the overall stability, output ripple voltage, and transient response of the DC-to-DC converter. The output ripple occurs due to variations in the charge stored in the output capacitor, the voltage drop due to the capacitor's ESR, and the voltage drop due to the capacitor's ESL. Calculate the output voltage ripple due to the output capacitance, ESR, and ESL as: VRIPPLE = VRIPPLE(C) + VRIPPLE(ESR) + VRIPPLE(ESL) where the output ripple due to output capacitance, ESR, and ESL is: VRIPPLE(C) = IP-P / (8 x COUT x fSW) VRIPPLE(ESR) = IP-P x ESR VRIPPLE(ESL) = (IP-P / tON) x ESL or (IP-P / tOFF) x ESL, whichever is greater and IP-P the peak-to-peak inductor current is: IP-P = [ (VIN - VOUT ) / fSW x L) ] x VOUT / VIN Use these equations for initial capacitor selection, but determine final values by testing a prototype or evaluation circuit. As a rule, a smaller ripple current results in less output voltage ripple. Since the inductor ripple current is a factor of the inductor value, the output voltage ripple decreases with larger inductance. Use ceramic capacitors for their low ESR and ESL at the switching frequency of the converter. The low ESL of ceramic capacitors makes ripple voltages negligible. Load transient response depends on the selected output capacitor. During a load transient, the output instantly changes by ESR x ILOAD. Before the controller can respond, the output deviates further, depending on the inductor and output capacitor values. After a short time (see the Load Transient Response graph in the
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Output Inductor Design
Use a 2H inductor with a minimum 2A-rated DC current for most applications. For best efficiency, use an inductor with a DC resistance of less than 20m and a saturation current greater than 3A (min). See Table 2 for recommended inductors and manufacturers. For most designs, derive a reasonable inductor value (LINIT) from the following equation: LINIT = VOUT x (VIN - VOUT) / (VIN x LIR x IOUT(MAX) x fSW) where fSW is the switching frequency (1MHz typ) of the oscillator. Keep the inductor current ripple percentage LIR between 20% and 40% of the maximum load current for the best compromise of cost, size, and performance. Calculate the maximum inductor current as: IL(MAX) = (1 + LIR / 2) x IOUT(MAX) Check the final values of the inductor with the output ripple voltage requirement. The output ripple voltage is given by: VRIPPLE = VOUT x (VIN - VOUT) x ESR / (VIN x LFINAL x fSW) where ESR is the equivalent series resistance of the output capacitors.
Input Capacitor Design
The input filter capacitor reduces peak currents drawn from the power source and reduces noise and voltage ripple on the input caused by the circuit's switching. The input capacitor must meet the ripple current requirement (IRMS) imposed by the switching currents defined by the following equation: IRMS = (1/ VIN ) x (IOUT2 x VOUT x (VIN - VOUT ))
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1MHz, All-Ceramic, 2.6V to 5.5V Input, 1.5A PWM Step-Down DC-to-DC Regulators MAX1951/MAX1952
Typical Operating Characteristics), the controller responds by regulating the output voltage back to its nominal state. The controller response time depends on the closed-loop bandwidth. A higher bandwidth yields a faster response time, thus preventing the output from deviating further from its regulating value. pole. C2 and R1 set a compensation zero. Calculate the dominant pole frequency as: fpEA = 1 / (2x CC x ROEA) Determine the compensation zero frequency is: fzEA = 1 / (2 x CC x RC) For best stability and response performance, set the closed-loop unity-gain frequency much higher than the modulator pole frequency. In addition, set the closedloop crossover unity-gain frequency less than, or equal to, 1/5 of the switching frequency. However, set the maximum zero crossing frequency to less than 1/3 of the zero frequency set by the output capacitance and its ESR when using POSCAP, SPCAP, OSCON, or other electrolytic capacitors.The loop-gain equation at the unity-gain frequency is: GEA(fc) x GMOD(fc) x VFB / VOUT = 1 where GEA(fc) = gmEA x R1, and GMOD(fc) = gmc x RLOAD x fpMOD/fC, where gmEA = 60S. R1 calculated as: R1 = VOUT x K / (gmEA x VFB x GMOD(fc)) where K is the correction factor due to the extra phase introduced by the current loop at high frequencies (>100kHz). K is related to the value of the output capacitance (see Table 1 for values of K vs. C). Set the error-amplifier compensation zero formed by R1 and C2 at the modulator pole frequency at maximum load. C2 is calculated as follows: C2 = (2 x VOUT x COUT / (R1 x IOUT(MAX)) As the load current decreases, the modulator pole also decreases; however, the modulator gain increases accordingly, resulting in a constant closed-loop unitygain frequency. Use the following numerical example to calculate R1 and C2 values of the typical application circuit of Figure 2a.
Compensation Design
The double pole formed by the inductor and output capacitor of most voltage-mode controllers introduces a large phase shift, which requires an elaborate compensation network to stabilize the control loop. The MAX1951/ MAX1952 utilize a current-mode control scheme that regulates the output voltage by forcing the required current through the external inductor, eliminating the double pole caused by the inductor and output capacitor, and greatly simplifying the compensation network. A simple type 1 compensation with single compensation resistor (R1) and compensation capacitor (C2) creates a stable and highbandwidth loop. An internal transconductance error amplifier compensates the control loop. Connect a series resistor and capacitor between COMP (the output of the error amplifier) and GND to form a pole-zero pair. The external inductor, internal current-sensing circuitry, output capacitor, and the external compensation circuit determine the loop system stability. Choose the inductor and output capacitor based on performance, size, and cost. Additionally, select the compensation resistor and capacitor to optimize control-loop stability. The component values shown in the typical application circuit (Figure 2) yield stable operation over a broad range of input-to-output voltages. The basic regulator loop consists of a power modulator, an output feedback divider, and an error amplifier. The power modulator has DC gain set by gmc x RLOAD, with a pole-zero pair set by RLOAD, the output capacitor (COUT), and its ESR. The following equations define the power modulator: Modulator gain: GMOD = VOUT / VCOMP = gmc x RLOAD Modulator pole frequency: fpMOD = 1 / (2 x x COUT x (RLOAD+ESR)) Modulator zero frequency: fzESR = 1 / (2 x x COUT x ESR) where, RLOAD = VOUT / IOUT(MAX), and gmc = 4.2S. The feedback divider has a gain of GFB = VFB / VOUT, where VFB is equal to 0.8V. The transconductance error amplifier has a DC gain, GEA(DC), of 70dB. The compensation capacitor, C2, and the output resistance of the error amplifier, R OEA (20M), set the dominant
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Table 1. K Value
DESCRIPTION COUT (F) 10 22 Values are for output inductance from 1.2H 0.55 to 2.2H. Do not use output inductors larger 0.47 than 2.2H. Use fC = 200kHz to calculate R1. K
VOUT = 1.5V IOUT(MAX) = 1.5A COUT = 10F RESR = 0.010 gmEA = 60S
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1MHz, All-Ceramic, 2.6V to 5.5V Input, 1.5A PWM Step-Down DC-to-DC Regulators
gmc = 4.2S fSWITCH = 1MHz RLOAD = VOUT / IOUT(MAX) = 1.5V / 1.5 A = 1 fpMOD = [1 / (2 x COUT x (RLOAD + RESR)] = [1 / (2 x x10 x10-6 x (1 + 0.01)] = 15.76kHz. fzESR = [1/(2 xCOUT RESR)] = [1 / (2 x x 10 x10-6 x 0.01)] = 1.59MHz. For 2H output inductor, pick the closed-loop unitygain crossover frequency (fC) at 200kHz. Determine the power modulator gain at fC: GMOD(fc) = gmc x RLOAD x fpMOD / fC = 4.2 x 1 x 15.76kHz / 200kHz = 0.33 then: R1 = VO x K / (gmEA x VFB x GMOD(fc)) = (1.5 x 0.55) / (60 x10-6 x 0.8 x 0.33) 51.1k (1%) C2 = (2 x VOUT x COUT) / (RC x IOUT(max) ) = (2 x 1.25 x 10 x 10-6) / (51.1k x 1.5) 209pF, choose 220pF, 10% particular attention. Follow these guidelines for good PC board layout: 1) Place decoupling capacitors as close to the IC as possible. Keep power ground plane (connected to PGND) and signal ground plane (connected to GND) separate. 2) Connect input and output capacitors to the power ground plane; connect all other capacitors to the signal ground plane. 3) Keep the high-current paths as short and wide as possible. Keep the path of switching current (C1 to IN and C1 to PGND) short. Avoid vias in the switching paths. 4) If possible, connect IN, LX, and PGND separately to a large copper area to help cool the IC to further improve efficiency and long-term reliability. 5) Ensure all feedback connections are short and direct. Place the feedback resistors as close to the IC as possible. 6) Route high-speed switching nodes away from sensitive analog areas (FB, COMP).
MAX1951/MAX1952
Applications Information
PC Board Layout Considerations
Careful PC board layout is critical to achieve clean and stable operation. The switching power stage requires
2.6V TO 5.5V IN C5 0.1F R1 51.1k C1 10F OFF C2 220pF ON Q1 R5 10k C3 0.1F R4 10 LX
L1 2H
1.5V AT 1.5A
MAX1951ESA
VCC COMP GND FB
R3 14.7k 1%
REF PGND R2 16.9k 1%
C4 10F
GND OPTIONAL SHUTDOWN CONTROL OUTPUT COMPONENT VALUES VOLTAGE (V) R1 (k) R2 (k) R3 (k) C2 (pF) 220 SHORT OPEN 0.8 33.2 220 14.7 16.9 1.5 51.1 220 30 14 2.5 82.5 220 75 24 3.3 110
Figure 2a. MAX1951 Adjustable Output Typical Application Circuit ______________________________________________________________________________________ 11
1MHz, All-Ceramic, 2.6V to 5.5V Input, 1.5A PWM Step-Down DC-to-DC Regulators MAX1951/MAX1952
2.6V TO 5.5V IN C5 0.1F R1 68k C1 10F OFF C2 220pF ON Q1 R5 10k R4 10 LX
L1 2H
1.8V AT 1.5A
MAX1952ESA-18
VCC COMP PGND FB
REF GND C3 0.1F
C4 10F
GND OPTIONAL SHUTDOWN CONTROL
Figure 2b. MAX1952 Fixed-Output Typical Application Circuit
12
______________________________________________________________________________________
1MHz, All-Ceramic, 2.6V to 5.5V Input, 1.5A PWM Step-Down DC-to-DC Regulators
Table 2. External Components List
COMPONENT (FIGURE 2) L1 FUNCTION Output inductor DESCRIPTION 2H 20% inductor Sumida CDRH4D28-1R8 or Toko A915AY-2R0M 10F 20%, 6.3V X5R capacitor Taiyo Yuden JMK316BJ106ML or TDK C3216X5R0J106MT 220pF 10%, 50V capacitor Murata GRM39X7R221K050AD or Taiyo Yuden UMK107CH221KZ 0.1F 20%, 16V X7R capacitor Taiyo Yuden EMK107BJ104MA, TDK C1608X7R1C104K, or Murata GRM 39X7R104K016AD 10F 20%, 6.3V X5R capacitor Taiyo Yuden JMK316BJ106ML or TDK C3216X5R0J106MT 0.1F 20%, 16V X7R capacitor Taiyo Yuden EMK107BJ104MA, TDK C1608X7R1C104K, or Murata GRM 39X7R104K016AD Figure 2a Figure 2a Figure 2a 10 5% resistor 10k 5% resistor NPN bipolar junction transistor Fairchild MMBT3904 Zetex FMMT413
MAX1951/MAX1952
C1
Input filtering capacitor
C2
Compensation capacitor
C3
Reference bypass capacitor
C4
Output filtering capacitor
C5
VCC bypass capacitor
R1 R2 R3 R4 R5 Q1
Loop compensation resistor Feedback resistor Feedback resistor Bypass resistor Shutdown transistor base current bias (optional) Shutdown transistor (optional)
Table 3. Component Suppliers
MANUFACTURER Murata Sumida Taiyo Yuden TDK Toko PHONE 650-964-6321 847-545-6700 800-348-2496 847-803-6100 1-800-PIK-TOKO FAX 650-964-8165 847-545-6720 847-925-0899 847-803-6296 408-943-9790
Chip Information
TRANSISTOR COUNT: 2500 PROCESS: BiCMOS
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13
1MHz, All-Ceramic, 2.6V to 5.5V Input, 1.5A PWM Step-Down DC-to-DC Regulators MAX1951/MAX1952
Package Information
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information go to www.maxim-ic.com/packages.)
SOICN .EPS
INCHES DIM A A1 B C e E H L MAX MIN 0.069 0.053 0.010 0.004 0.014 0.019 0.007 0.010 0.050 BSC 0.150 0.157 0.228 0.244 0.016 0.050
MILLIMETERS MAX MIN 1.35 1.75 0.10 0.25 0.35 0.49 0.19 0.25 1.27 BSC 3.80 4.00 5.80 6.20 0.40 1.27
N
E
H
VARIATIONS:
1
INCHES
MILLIMETERS MIN 4.80 8.55 9.80 MAX 5.00 8.75 10.00 N MS012 8 AA 14 AB 16 AC
TOP VIEW
DIM D D D
MIN 0.189 0.337 0.386
MAX 0.197 0.344 0.394
D C
A e B A1
0 -8 L
FRONT VIEW
SIDE VIEW
PROPRIETARY INFORMATION TITLE:
PACKAGE OUTLINE, .150" SOIC
APPROVAL DOCUMENT CONTROL NO. REV.
21-0041
B
1 1
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
14 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 (c) 2002 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products.

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